FIG. 12 is a block diagram showing the general structure of a conventional switching power supply.
Various inverters and converters are used for the switching power supply. Hereinafter, the conventional switching power supply will be described in connection with a DC-DC converter as a typical example. In the conventional switching power supply shown in FIG. 12, the output voltage VOUT from DC-DC converter 100 is fed back to converter control section 200. Converter control section 200 compares the output voltage VOUT with a reference value, generates a PWM signal (VCONT) based on the comparison results and outputs the PWM signal (VCONT). DC-DC converter 100 switches on and off a semiconductor switch such as a MOSFET and a bipolar transistor or a mechanical switch such as a relay based on the PWM signal (VCONT) outputted to set the output voltage VOUT therefrom to be close to the reference value.
Concrete examples of the DC-DC converter in FIG. 12 are illustrated in FIGS. 13 through 20. FIG. 13 shows a first buck converter used as the DC-DC converter in FIG. 12. FIG. 14 shows a second buck converter used as the DC-DC converter in FIG. 12. FIG. 15 shows a first boost converter used as the DC-DC converter in FIG. 12. FIG. 16 shows a second boost converter used as the DC-DC converter in FIG. 12. FIG. 17 shows a first buck-boost converter used as the DC-DC converter in FIG. 12. FIG. 18 shows a second buck-boost converter used as the DC-DC converter in FIG. 12. FIG. 19 shows a first flyback converter used as the DC-DC converter in FIG. 12. FIG. 20 shows a second flyback converter used as the DC-DC converter in FIG. 12.
Now the conventional DC-DC converters will be described briefly below. The output voltage Vo (VOUT in FIG. 13) from the conventional buck converter shown in FIG. 13 is expressed as Vo=Ton/(Ton+Toff)×Vi using the ON-period (conducting period) Ton of main switch S1 (104), the OFF-period (non-conducting period) Toff thereof, and the input voltage Vi (VIN in FIG. 13). When the input voltage Vi varies, the DC-DC converter shown in FIG. 13 adjusts the ratio of the ON-period Ton and the OFF-period Toff of main switch S1 (104) to absorb the change caused in the input voltage Vi and to keep the output voltage Vo at a certain value. The term Ton/(Ton+Toff) in the above-described equation is an ON-period ratio D. Using the ON-period ratio D, the above-described equation is given by Vo=D ×Vi. The above-described equation does not consider any loss. The above-described equation describes the operation of the DC-DC converter in the continuous mode, in which a current flows through coil 120 continuously. (In the following, the DC-DC converters will be described similarly in connection with the continuous operation mode thereof.) The conventional buck converter shown in FIG. 14 is a modification of the first buck converter shown in FIG. 13. In FIG. 14, subsidiary switch S2 (106) is disposed in substitution for flywheel diode D2 (107) shown in FIG. 13. Since the DC-DC converter shown in FIG. 14 works essentially in the same manner as the DC-DC converter shown in FIG. 13, the operation of the DC-DC converter shown in FIG. 14 is not described herein. Subsidiary switch S2 (106) is switched on and off oppositely to the switching on and off of main switch S1 (104).
The first boost converter shown in FIG. 15 is a DC-DC converter, the output voltage Vo therefrom is expressed by Vo=((Ton+Toff)/Toff)×Vi=(1/ (1−D))×Vi. The first boost converter shown in FIG. 15 superimposes the energy stored in coil 120 while main switch S1 (104) is conductive (during Ton) and the voltage Vo thereof is given by ((Ton+Toff)/Toff)×Vi=(1/(1−D))×Vi, onto the input energy, the voltage thereof is Vin, while main switch S1 (104) is switched off (during Toff). The conventional boost converter shown in FIG. 16 is a modification of the first boost converter shown in FIG. 15. In FIG. 16, subsidiary switch S2 (106) is disposed in substitution for flywheel diode D2 (107) shown in FIG. 15. Since the DC-DC converter shown in FIG. 16 works essentially in the same manner as the DC-DC converter shown in FIG. 15, the operation of the DC-DC converter shown in FIG. 16 is not described herein. Subsidiary switch S2 (106) is switched on and off oppositely to the switching on and off of main switch S1 (104).
The first buck-boost converter shown in FIG. 17 is a DC-DC converter, the output voltage Vo therefrom is expressed by Vo=−Ton/Toff×Vi=−(D/(1−D))×Vi. The first buck-boost converter uses the energy stored in coil 120 while main switch S1 (104) is conductive (during Ton) for making a current flow through coil 120, while main switch S1 (104) is switched off (during Toff), to the direction same with the direction, to which a current flows through coil 120 while main switch S1 (104) is conductive (during Ton). The DC-DC converter shown in FIG. 17 facilitates setting the output voltage Vo not only to be higher than the input voltage Vi, but also to be lower than the input voltage Vi. The conventional buck-boost converter shown in FIG. 18 is a modification of the first buck-boost converter shown in FIG. 17. In FIG. 18, subsidiary switch S2 (106) is disposed in substitution for flywheel diode D2 (107) shown in FIG. 17. Since the DC-DC converter shown in FIG. 18 works essentially in the same manner as the DC-DC converter shown in FIG. 17, the operation of the DC-DC converter shown in FIG. 18 is not described herein. Subsidiary switch S2 (106) is switched on and off oppositely to the switching on and off of main switch S1 (104).
The first flyback converter shown in FIG. 19 is a DC-DC converter, the output voltage Vo therefrom is expressed by Vo=(N2/N1)×(Ton/Toff)×Vi=(N2/N1)×(D/(1−D))×Vi. The first flyback converter stores an energy in transformer 125 through primary winding N1 thereof, while main switch S1 (104) is conductive (during Ton). As main switch S1 (104) is switched off (during Toff), the first flyback converter outputs the energy stored in transformer 125 to output capacitor 130 via secondary winding N2 in transformer 125. Herein, the winding ratio of transformer 125 is given by N2/N1. The conventional flyback converter shown in FIG. 20 is a modification of the first flyback converter shown in FIG. 19. In FIG. 20, subsidiary switch S2 (106) is disposed in substitution for flywheel diode D2 (107) shown in FIG. 19. Since the DC-DC converter shown in FIG. 20 works essentially in the same manner as the DC-DC converter shown in FIG. 19, the operation of the DC-DC converter shown in FIG. 20 is not described herein. Subsidiary switch S2 (106) is switched on and off oppositely to the switching on and off of main switch S1 (104).
The switches S1 and S2 in FIGS. 14, 16, 18 and 20 are switched on and off based on the PWM signal (VCONT) via driver circuit 102. While the switch S1 is ON (conductive), the subsidiary switch S2 is forced to be OFF (non-conductive). While the switch S1 is OFF (non-conductive), the subsidiary switch S2 is forced to be ON (conductive). In the switching power supply apparatuses shown in FIGS. 13 through 20, a discontinuous current mode is caused when the load is light. In the discontinuous current mode, both the main switch S1 and the diode D2 are not conductive or both the switches S1 and S2 are OFF. The switching power supply apparatuses shown in FIGS. 13 through 20 control the output voltage thereof by adjusting the ON-period ratio D even in the discontinuous current mode. (Detailed description on the discontinuous current mode is not made herein.)
Now the operation of converter control section 200 in FIG. 12 will be described more in detail below with reference to FIG. 21 that schematically describes the operation waveforms of conventional converter control section 200 and FIG. 22 that shows the structural example thereof.
Converter control section 200 in FIG. 12 includes detector circuit 210, reference voltage supply VREF (220), and control circuit 230. Control circuit 230 includes error amplifier circuit 232, comparator circuit 234 and oscillator circuit 236. Comparator circuit 234 outputs a PWM signal (VCONT). Oscillator circuit 236 can output a triangular wave, saw-tooth wave, sinusoidal wave and such various waves. Herein, the operation of converter control section 200 is described in connection with a widely-used triangular wave that oscillator circuit 236 outputs. Therefore, oscillator circuit 236 is a triangular-wave generator herein. The upper waveforms in FIG. 21 typically describe the comparison of an error voltage VE and a triangular wave VOSC. The lower waveform in FIG. 21 describes an output signal VCONT (PWM signal) generated based on the waveform comparison illustrated by the upper waveforms in FIG. 21. In FIG. 21, TS is the period of the output waveform VOSC from oscillator circuit 236 and equal to the period (switching period), for which main switch S1 (104) in the switching power supply is switched on and off. In FIG. 21, t1 is the ON-period (above-described Ton) of main switch S1 (104) and t2 is the OFF-period (above-described Toff) of main switch S1 (104). The ON-period ratio D of main switch S1 (104) is given by D=t1/TS. The period TS of the output waveform VOSC is given by the following formula.TS=t1+t2=Ton+Toff
The converter control section shown in FIG. 22 employs detecting resistors R1 and R2 for detector circuit 210 and an operational amplifier (error amplifier in the figure) for error amplifier circuit 232. The converter control section shown in FIG. 22 also employs a triangular-wave generator for oscillator circuit 236 and a comparator (PWM comparator in the figure) for comparator circuit 234. Alternatively, detector circuit 210 is not employed and the output voltage VOUT is set to be equal to the voltage V0. Still alternatively, a reference value is applied from the outside in substitution for reference voltage supply VREF (220) and the applied reference value is changed with time. In the alternative cases, the comparator control section operates essentially in the same manner as in the case described at first in this paragraph. Therefore, the operation of conventional converter control section 200 will be described below in connection with the circuit configuration shown in FIG. 22.
FIG. 23A is a block functional diagram describing the signal processing process of comparator circuit (comparator) 234 in the conventional converter control section shown in FIG. 22. FIGS. 23B through 23D are wave charts describing the operation waveforms of comparator circuit (comparator) 234 in the conventional converter control section shown in FIG. 22.
Referring now to FIG. 23A, the contents of the signal processing conducted by comparator 234 may be divided generally into input stage 2341, amplifying stage 2342 and output stage 2343. Comparator 234 generates the output signal VCONT (PWM signal) in output stage 2343 based on the input waveforms blunted in input stage 2341 and amplifying stage 2342 and outputs the output signal VCONT (PWM signal).
The waveforms inputted to input stage 2341 are shown in the upper parts of FIGS. 23B through 23D. The waveforms inputted to amplifying stage 2342 are shown in the middle parts of these figures. The waveforms outputted from output stage 2343 are shown in the lower parts of these figures. Usually, the delay td1 is caused on the waveform outputted from output stage 2343 mainly by the blunting caused on the input voltage V1 inputted to amplifying stage 2342. Since the input voltage V1 inputted to amplifying stage 2342 changes greatly across the threshold of amplifying stage 2342 in the state shown in FIG. 23B, the output signal VCONT from output stage 2343 is generated in the form of pulses. More detailed explanations will be made below. In the state shown in FIG. 23B, the error signal VE outputted from error amplifier 232 is compared in PWM comparator 234 with the triangular wave VOSC outputted from triangular wave generator 236. When the error signal VE is equal to VE1 and VE1 is higher than VOSC, the output signal VCONT from PWM comparator 234 is HIGH. As the triangular wave VOSC rises such that VOSC is higher than VE1, the output signal VCONT shifts to the LOW-state after the response delay time td1 caused in the shifting to the LOW-state of PWM comparator 234 elapses. As the triangular wave VOSC lowers such that VE1 is higher than VOSC, the output signal VCONT shifts to the HIGH-state after the response delay time td2 caused in the shifting to the HIGH-state of PWM comparator 234 elapses. As described above, the output signal VCONT repeats shifting to the LOW-state and the HIGH-state alternately after the delay for the response delay times td1 and td2 necessary for the shifting (output signal shift).
In the state shown in FIG. 23C, the error signal VE and the triangular wave VOSC reverse the height relation thereof in input stage 2341 while the changing input voltage V1 inputted to amplifying stage 2342 is still close to (not departing far from) the threshold of amplifying stage 2342. Therefore, the LOW-period of the output signal VCONT from output stage 2343 becomes short. More detailed explanations will be made below. When the error signal VE is higher than VE1 shown in FIG. 23B but slightly lower than VE2 shown in FIG. 23D in the state shown in FIG. 23C, the output signal VCONT shifts to the LOW-state after the response delay time td1 caused in the shifting to the LOW-state in PWM comparator 234 elapses. However, the error signal VE and the triangular wave VOSC reverse the height relation thereof in input stage 2341 soon. Therefore, the output signal VCONT shifts to HIGH-state after a short response delay time td2 caused in the shifting to the HIGH-state in PWM comparator 234 elapses. Therefore, the LOW-period of the output signal VCONT becomes short.
In the state shown in FIG. 23D, the error signal VE and the triangular wave VOSC reverse the height relation thereof in input stage 2341 before the lowering input voltage V1 reaches the threshold of amplifying stage 2342. Therefore, the input voltage V1 rises before reaching the threshold of amplifying stage 2342 and the output signal VCONT keeps the HIGH-state thereof without shifting to the LOW-state. More detailed explanations will be made below. In the state shown in FIG. 23D, the error signal VE becomes higher than the state thereof shown in FIG. 23C and equal to VE2. In other words, as the time ta in FIG. 23D becomes almost equal to the response delay time td1 caused in the shifting to the LOW-state in PWM comparator 234, PWM comparator 234 cannot respond, setting the output signal VCONT always to be HIGH. In the state in which PWM comparator 234 cannot respond, it is implied that the response delay time td2 caused in the shifting to the HIGH-state in PWM comparator 234 is shortened rapidly, causing zero delay time td2 finally.
In the above, the description is made in connection with the case, in which the error signal VE rises gradually. However, as the error signal VE lowers gradually and the time TS-ta becomes almost equal to the response delay time td2 caused in the shifting to the HIGH-state in PWM comparator 234, PWM comparator. 234 cannot respond, setting the output signal VCONT always at the LOW-state. In the state in which PWM comparator 234 cannot respond, it is implied that the response delay time td1 caused in the shifting to the LOW-state in PWM comparator 234 is shortened rapidly, causing zero delay time td1 finally.
FIG. 24 is a graph relating the ON-period ratio D in the conventional converter control section shown in FIG. 22 with the error signal VE.
As shown in FIG. 24, the conventional converter control section cannot output any pulse in the ON-period ratio range D smaller than td2/TS or higher than (TS−td1)/TS. (Here, td1 and td2 are the td1 and td2 shown in FIG. 23B.) In the D ranges described above, a region, in which any pulse with the pertinent ON-period ratio D is not generated suddenly, is caused as described above. Due to this, the span, for which the ON-period ratio D changes linearly, is narrowed.
The response delay caused by the PWM comparator used in the conventional converter control section is from several tens ns to several hundreds ns, generally. To meet the recent demands for down-sizing the switching power supply apparatuses, the switching frequency has been increased. Especially, the switching frequency of some DC-DC converters reaches several MHz. If the response delay of the PWM comparator is 100 ns, the ON-period ratio D controllable by the conventional converter control section will be 10 to 90%, when the switching frequency is 1 MHz. If the switching frequency is 10 MHz, the response time of the PWM comparator will be equal to the switching period, making it impossible to conduct switching any more. Therefore, the conventional converter control section prevents the switching power supply from increasing the switching frequency thereof.
For increasing the switching frequency of the switching power supply, Unexamined Japanese Patent Application Publication No. 2002-261588 and Unexamined Japanese Patent Application Publication No. 2004-282352 disclose circuits for generating a triangular wave or a saw-tooth wave causing neither overshooting nor undershooting at high frequencies. Since the disclosed circuits generate a triangular wave or a saw-tooth wave stably at high frequencies, the disclosed circuits provide techniques for precisely controlling the ON-period ratio or the OFF-period ratio using the triangular wave or the saw-tooth wave. The following Unexamined Japanese Patent Application Publication No. 2005-143197 discloses a DC-DC converter that facilitates obviating the problems caused by narrow-width-pulse propagation in generating a signal for determining the maximum ON-period ratio or the maximum OFF-period ratio and controlling the maximum ON-period ratio or the maximum OFF-period ratio very precisely even at a high frequency.
As described above, it is impossible to control the ON-period ratio precisely due to the response delay of the PWM comparator when the ON-period ratio is very large or very small. The three above-mentioned Japanese Patent documents do not describe anything on the above-described problem nor disclose any countermeasures for solving the above-described problem.
In view of the foregoing, it would be desirable to provide a control circuit and a control method for controlling a switching power supply that facilitate greatly widening the HIGH-period ratio range or the LOW-period ratio range of the PMW signal in regulating the output from the switching power supply by adjusting the ON-period ratio or the OFF-period ratio of the switching device in the switching power supply by the PWM method.